(1) Field of the Invention
The present invention relates to a receiving device to be used for a spread spectrum radar apparatus using a spread spectrum scheme, and particularly to a receiving device that can obtain a radar spectrum that is precise and has stable intensity without depending on a phase of a reception signal.
(2) Description of the Related Arts
In recent years, vehicles have been equipped with radar apparatuses for use in detecting objects, such as vehicles ahead and obstacles behind the vehicles. There are high expectations for such radar apparatuses in terms of improvements in safety, such as avoiding collision, and in convenience of drivers as represented by support for reverse driving operations. Thus, technical developments in radar apparatus to be mounted in vehicles (hereinafter referred to as in-vehicle radar apparatuses) have been active. One of the important tasks of in-vehicle radar apparatuses is to suppress the influence of radio waves to be emitted from radar apparatuses that are mounted in other vehicles and are similar to the radar apparatus of the own vehicle. As an example, radar apparatuses using a spread spectrum scheme (hereinafter referred to as spread spectrum radar apparatuses) have been suggested.
The spread spectrum radar apparatuses modulate emitted radio waves using pseudo-noise codes (hereinafter referred to as PN codes) for spreading waves. Receivers for such spread spectrum radar apparatuses despread waves reflected from objects, using the same PN codes that have been used in modulating the emitted radio waves. Thus, either radio waves that have been modulated using different codes or radio waves that have been emitted from radar apparatuses that use other schemes with no code modulation are suppressed in the receivers. Furthermore, since the emitted radio waves are frequency-spread using the PN codes, electric power on a frequency unit basis and the impact to other wireless systems can be reduced. Furthermore, a relationship between a distance resolution and a maximum detectable range may be freely set by adjusting chip rates and code cycles of the PN codes. Furthermore, the peak power never becomes large due to continuous transmission of radio waves.
FIG. 1 illustrates a configuration of a receiving unit 300 included in a receiving device of a conventional spread spectrum radar apparatus disclosed in Unexamined Japanese Patent Application Publication No. 2005-72735 , published Mar. 17, 2005 (hereinafter referred to as Patent Reference 1). The receiving unit 300 includes a receiving antenna 301, a low noise amplifier 302, a despreading unit 303, a phase shifter 304, a quadrature demodulator 305, and buffer amplifiers 307a and 307b. 
A transmission signal that has been spread over a wide band and then transmitted by a transmitting device is reflected from an object at a certain distance. The signal reflected from the object is received by the receiving antenna 301 of the receiving unit 300 in FIG. 1. The despreading unit 303 despreads the reflected signal using a PN code provided from a reception PN code generating unit 310 to the despreading unit 303, and converts the despread signal into a narrow-band signal. Then, the narrow-band signal is separated into 2 different narrow-band signals that pass through differential transmission lines that have a phase difference by 180 degrees. The 2 narrow-band signals are down-converted by balanced modulators 305a and 305b based on 2 different local oscillator signals in order to generate an in-phase signal and a quadrature signal respectively from the balanced modulators 305a and 305b. Here, the 2 different local oscillator signals have a phase difference by approximately 90 degrees and are generated by a local oscillator 306 and a phase shifter 304, respectively. A sum of squares of the in-phase signal and the quadrature signal is calculated to obtain intensity of the signals. Furthermore, a control unit (not illustrated) included in the receiving unit 300 controls how long a PN code identical to the PN code used in the transmitting device is delayed for use in the receiving unit 300. Furthermore, the signal processing unit 320 calculates a distance to the object by performing signal processing on the signal received by the receiving unit 300 so that the distance may be reflected to a radar spectrum.
FIG. 2 illustrates a circuit configuration, disclosed in Patent Reference 1, including the despreading unit 303 of the receiving unit 300 and the balanced modulators 305a and 305b included in the quadrature demodulator 305. The despreading unit 303 and the balanced modulators 305a and 305b are double balanced I/O switching circuits, and transistors included in amplifiers of Gilbert cell mixer circuits are omitted in FIG. 2. A current source circuit 331 can be commonly used by integrating the despreading unit 303, and the balanced modulators 305a and 305b into a circuit. Thereby, a current value of a current to be consumed can be reduced, and electric power consumption can also be reduced. Furthermore, reception signals provided from collectors of transistors included in the despreading unit 303 can be directly received by emitters of transistors included in the balanced modulators 305a and 305b. Thus, distortion effect can be suppressed and further, a size of a chip can be reduced.
FIG. 3 simply illustrates operations of the receiving unit 300 illustrated in FIG. 2. A reception signal is converted from an unbalanced signal into a balanced signal by a balun 330. As illustrated in FIG. 3, currents A and B are carried by balanced output transmission lines of the balun 330. Furthermore, transistors Q1, Q7, Q9, Q12, Q13, and Q16 are turned on, and other transistors are turned off.
The current A flows through the transistor Q1, and the current B flows through the transistor Q7. Since the transistors Q9, Q12, Q13, and Q16 are turned on in the quadrature demodulator 305, the current A is made up of a current A1 that flows through the transistor Q9 and a current A2 that flows through the transistor Q13. The current B is made up of a current B1 that flows through the transistor Q12 and a current B2 that flows through the transistor Q16. Here, when the current A1 is equal to the current A2, intensity of a signal transmitted from a terminal OUT1 is equal to that of a signal transmitted from a terminal OUT3. When the current B1 is equal to the current B2, intensity of a signal transmitted from a terminal OUT2 is equal to that of a signal transmitted from a terminal OUT4. In other words, an in-phase balanced signal made up of the signals transmitted from the terminals OUT1 and OUT2 is equal to a quadrature balanced signal made up of the signals transmitted from the terminals OUT3 and OUT4. Hereinafter, processing for obtaining a radar spectrum is described more specifically using equations.
Equation 1 expresses 2 balanced signals RF1 and RF2 obtained through the despreading unit 303, and Equation 2 expresses local oscillator signals LO_I and LO_Q that have a phase difference by 90 degrees and that are provided by the quadrature demodulator 305.
P1 and P2 represent intensities of the balanced signals RF1 and RF2, and φ represents a phase of a reception signal RF. The reception signal RF is demodulated by the quadrature demodulator 305 based on the local oscillator signals LO_I and LO_Q.RF1=P1 cos(ω1t+φ)   (Equation 1)RF2=P2 cos(ω1t+φ+π)   (Equation 2)LO_I=cos ω2tLO_Q=sin ω2t
Equation 3 expresses an output in-phase signal IF_I and an output quadrature signal IF_Q obtained by removing a signal component having larger frequency through a filter from the demodulated reception signal RF.IF—I=(P1/2)cos {(ω1−ω2)t+φ}  (Equation 3)IF—Q=(P2/2)sin {(ω1−ω2)t+φ}
Equation 4 expresses a sum of squares T of the in-phase signal IF_I and the quadrature signal IF_Q.T=√[(P1/2)2 cos2{(ω1−ω2)t+φ}+(P2/2)2 sin2{(ω1−ω2)t+φ}]  (Equation 4)
The sum of squares T reflects a peak of a radar spectrum. When the 2 balanced signals RF1 and RF2 obtained through the despreading unit 303 have different values representing the intensities P1 and P2, the sum of squares T of the in-phase signal IF_I and the quadrature signal IF_Q varies according to a phase φ of the reception signal RF. However, when the intensity P1 is equal to that of P2, the sum of squares T of the in-phase signal IF_I and the quadrature signal IF_Q becomes a constant value.
Here, the intensities P1 and P2 of the 2 balanced signals RF1 and RF2 obtained through the despreading unit 303 depend on a difference between current values of the currents A1 and B1 or a difference between current values of the currents A2 and B2. When absolute values of these two differences of the current values become equal, according to a result of Equation 4, the sum of squares T of the in-phase signal IF_I and the quadrature signal IF_Q becomes a constant value and can secure intensity of the reception signal RF at its peak.